Satisfying consumer demand for mobile handset applications, like streaming video, mobile e-mail, web browsing, etc., requires mobile telecommunications systems that support high data rates. Unfortunately, the radio frequency (RF) spectrum is a limited and highly regulated resource, which makes developing mobile telecommunications systems that support these data-intensive applications tremendously challenging. To increase data throughput in a given amount of RF spectrum, various techniques have been, and continue to be, developed, including modulation schemes that are more spectrally efficient and communication protocols that allow mobile handsets and base stations to communicate with reduced latency.
In second generation (2G) Global System for Mobile Communications (GSM) telecommunications systems, data rates are increased by using either the General Packet Radio Service (GPRS) or the Enhanced Data Rates for GSM Evolution (EDGE) service (often referred to as 2.5G and 2.75G technologies). GSM, GPRS and EDGE utilize time division multiple access (TDMA), a channel access method that allows several users to share the same frequency channel. Transmitters transmit in rapid succession, one burst after the other, each according to pre-assigned time slots of a TDMA frame. This allows each transmitter to share the same radio frequency channel while using only part of the channel's total capacity. GSM supports data rates of up to only 14.4 kilobits per second (Kbps), allowing only one data slot per TDMA frame. GPRS enhances data throughput (up to a theoretical limit of 115 Kbps) by allowing eight data slots per TDMA frame. EDGE further enhances data throughput (up to a theoretical limit of 474 Kbps) by also utilizing eight data slots per TDMA frame and further employing a higher order modulation scheme (8 phase shift keying (8PSK)).
Third generation (3G) systems, such as the Universal Mobile Telecommunications System (UMTS) and the Freedom of Mobile Multimedia Access (FOMA) system deployed in Japan, achieve increased data rates by using the Wideband Code Division Multiple Access (W-CDMA) air interface. W-CDMA employs a 5 MHz channel bandwidth, compared to the 200 KHz channel bandwidth used in GSM systems. The increased channel bandwidth, together with a more spectrally efficient non-constant envelope modulation scheme, allows data rates as high as 2 megabits per second (Mbps) to be achieved.
Data throughput can be further increased in 3G systems by employing High Speed Uplink Packet Access (HSUPA) or High Speed Downlink Packet Access (HSDPA). HSDPA employs adaptive modulation and coding (AMC)—a technique that utilizes one of two different non-constant envelope modulation schemes depending on the quality of the radio link between a mobile handset and a base station. When the radio link is poor, QPSK modulation scheme is used. However, when the radio link is good, 16 quadrature amplitude modulation (16QAM) is used. 16QAM is a higher order modulation scheme than QPSK, and its use nearly doubles the data rate over that achievable using QPSK. Data rates are also increased by fast packet scheduling at the base station and fast retransmissions from the base station, known as hybrid automatic repeat-request (HARQ). Similar techniques are used to increase data rates in HSUPA. By using AMC and the refined handset to base station protocols, a theoretical downlink (base station to handset) performance of 14.4 Mbps can be achieved using HSDPA, while a theoretical uplink performance of 5.76 Mbps can be achieved using HSUPA. These data rates rival or even exceed land-based digital subscriber line (DSL) technology, and allow broadband Internet and video calling capabilities.
Although the non-constant envelope modulation schemes used in 3G systems are more spectrally efficient than constant envelope modulation schemes, their use introduces a number of problems. One especially significant problem relates to energy efficiency. In conventional quadrature-modulator-based transmitters power must be backed off in order to prevent clipping of the signal peaks of non-constant envelope signals. Power back-off is achieved by biasing the transmitter's power amplifier (PA) so that the PA is forced to operate in its linear region of operation, over the full range of output powers the PA must be configured to operate. In a typical design, this is achieved by biasing the PA so that its peak output power does not exceed the PA's 1 dB compression point, which defines the input power at which the gain of the PA drops by 1 dB from its ideal, linear response value. The degree of power back-off required depends on the peak-to-average ratio (PAR), which in turn depends on the particular non-constant envelope modulation scheme being used. The higher the PAR, the more the output power of the PA must be backed off. For example, for a PAR of 3 dB, an average power of 10 dBm, and a peak output power of 13 dBm, a linear PA response requires that the average output power of the PA be backed off by at least 3 dB in order for the peak output power not to exceed the 1 dB compression point.
Employing power back-off does help to ensure PA linearity. However, it also undesirably results in a significant reduction in energy efficiency. The energy efficiency of a transmitter is determined in large part by how efficient the transmitter's PA is. The efficiency of a PA is defined as the ratio of the PA RF output power to the direct current (DC) power supplied to the PA. Efficiency is therefore high when the PA is operating at a high RF output power, but low when the PA is operating at low RF output powers. In most any practical application, the PA operates at high or peak RF output powers only for very short periods of time. For all other times, the RF output power is backed off. Consequently, power back-off results in a substantial reduction in energy efficiency.
The low energy efficiency of conventional quadrature-modulator-based transmitters is a major problem, particularly in a mobile handset since the handset's transmitter and PA are battery-powered. Fortunately, a more efficient type of communications transmitter known as a polar transmitter is available. As explained below, in a polar transmitter the amplitude information (i.e., the signal envelope) is temporarily removed from the non-constant envelope signal. This affords the ability to operate the polar transmitter's PA in its nonlinear region, where it is more efficient at converting energy from the transmitter's power supply into RF power than it is when configured to operate in its linear region.
FIG. 1 is a drawing showing the basic elements of a typical polar transmitter 100. The polar transmitter 100 comprises a symbol mapper 102; an oversampling block 104; a pulse shaping filter 106; a Coordinate Rotation Digital Computer (CORDIC) converter 108; an amplitude path including an amplitude path digital-to-analog converter (DAC) 110 and an envelope modulator 112; a phase path including a phase path DAC 114 and a phase modulator 116; an RF PA 118; and an antenna 120. Collectively, the symbol mapper 102, oversampling block 104 and pulse shaping filter 106 form a baseband modulator 101. The symbol mapper 102 operates to map data bits in a digital message to be transmitted to a signal constellation defined by an applicable modulation scheme, according to a symbol clock, to produce an in-phase sequence of symbols I(n) and a quadrature phase sequence of symbols Q(n). The oversampling block 104 increases the data rate of the I(n) and Q(n) sequences of symbols by a rate determined by an oversampling clock, to increase resolution and eliminate the need for (or at least relax the requirements of) subsequent reconstruction filters (not shown). The pulse shaping filter 106 (e.g., a root-raised-cosine filter) pulse shapes the oversampled I and Q sequences of symbols to band-limit the signals and reduce inter-symbol interference.
The digital pulse-shaped I(t) and Q(t) signals produced at the output of the baseband modulator 101 are converted from rectangular to polar coordinates by the CORDIC converter 108. The amplitude and phase path DACs 110 and 114 convert the resulting amplitude and phase component signals ρ(t) and θ(t) to analog amplitude and phase modulation signals. In the amplitude path, the envelope modulator 112 modulates a direct current power supply voltage Vsupply (e.g., as provided by a battery) according to the amplitude information in the amplitude modulation signal. The resulting amplitude-modulated power supply signal Vs(t) is coupled to the power supply port of the PA 118. In the phase path, the phase modulator 116 operates to modulate an RF carrier signal according to the phase information in the phase modulation signal, to produce a phase-modulated RF carrier signal which is coupled to the RF input port RFIN of the PA 118.
Because the phase-modulated RF carrier signal has a constant envelope, the PA 118 can be operated in its nonlinear region of operation without the risk of signal peak clipping. Typically, the PA 118 is implemented as a switch-mode PA operating between compressed and cut-off states. When configured in this manner, the envelope information in the amplitude-modulated power supply signal Vs(t) is restored at the RF output RFOUT of the PA 118, as the PA 118 amplifies the phase-modulated RF carrier signal. The desired amplitude- and phase-modulated RF carrier signal appearing at the RF output RFOUT of the PA 118 is coupled to the antenna 120, which finally radiates the signal over the air to a remote receiver (e.g., a base station).
While polar transmitters are able to process and transmit non-constant envelope signals at higher efficiencies than can be realized in more conventional quadrature-modulator-based transmitters, the amplitude and phase component signals ρ(t) and θ(t) typically have higher signal bandwidths compared to the rectangular-coordinate I(t) and Q(t) signals. This so-called “bandwidth expansion” phenomenon occurs during the rectangular-to-polar conversion process performed by the CORDIC converter 108, resulting in amplitude and phase component signals ρ(t) and θ(t) containing high-frequency events. These high-frequency events are undesirable, since they can cause the transmission spectrum to extend beyond the limits of the intended band-limited channel, resulting in adjacent channel interference and an increase in receive band (Rx band) noise.
High-frequency events also have an impact on modulation accuracy. Accurate representation of high-frequency events necessitates a high signal processing rate. Not only does this result in increased energy consumption, in situations where the required signal processing rate exceeds the capabilities of the underlying hardware substantial in-band signal distortion occurs.
When high-frequency events are present in the amplitude component signal ρ(t), the envelope modulator 112 is unable to accurately track the signal envelope. One reason for this is that the envelope modulator 112 is usually implemented as a switch-mode power supply containing power transistors that are operated as switches. Due to their large gate capacitances, these power transistors have a limited switching speed. This constrains the bandwidth handling capability of the envelope modulator 112. Accordingly, high-frequency events in the amplitude component signal ρ(t) that are beyond the bandwidth handling capability of the envelope modulator 112 result in degraded modulation accuracy.
High-frequency events in the phase component signal θ(t) that exceed the bandwidth handling capability of the phase modulator 116 also degrade modulation accuracy. In a digital polar transmitter implementation, the phase is carried through the phase path of the polar transmitter 100 as the phase difference Δθ between sample clocks. A voltage-controlled oscillator (VCO) within the phase modulator 116 integrates the phase-difference Δθ in successive samples to produce the desired phase-modulated RF carrier signal. The phase-difference Δθ determines how fast the VCO must integrate. When high-frequency events are present in the phase component signal θ(t), the phase-difference Δθ between sample clocks can be large enough that it exceeds the bandwidth handling capability of the VCO. Under these conditions, the signal phase of the phase-modulated RF carrier signal ends up lagging or leading the desired signal phase and phase accumulation error results. The phase accumulation error manifests itself as phase jitter at the output of the phase modulator 116 and results in degraded modulation accuracy.
Various techniques have been proposed to mitigate the effects of high-frequency events in polar domain signals. One commonly-used approach, known as “hole blowing,” operates on the assumption that low amplitude I-Q samples correlate in time with high-frequency events in the polar domain. The rationale for this assumption is illustrated in FIG. 2, which is a simplified drawing of a signal trajectory of a non-constant envelope signal in the quadrature domain. As shown, as the signal traverses closer and closer to the origin, the magnitude of the signal approaches zero. When the signal eventually passes through the origin, a near 180° phase change occurs. This abrupt change in phase is indicative of a high-frequency event. To prevent this from occurring, hole blowing algorithms operate to condition low-magnitude samples that fall below a predetermined low-magnitude threshold α, so that the conditioned samples fall on a point on a circle (i.e., hole) centered at the origin and having a radius equal to the predetermined low-magnitude threshold α (see FIG. 3). By doing this, the signal trajectory is altered so that the signal does not pass too close to the origin.
While hole blowing can be an effective in some applications, it can be detrimental in others. The problem relates to the assumption that low-magnitude events always provide an indication of high-frequency content. This is not necessarily the case. For example, as illustrated in FIG. 4, while low-magnitude events 1 and 2 in the amplitude component signal ρ(t) are seen to correspond to high-frequency events 1 and 2 in the phase-difference component signal Δθ, low-magnitude event 3 in the amplitude component signal ρ(t) does not correspond to any high-frequency event in the phase-difference component signal Δθ. Such a condition may arise for signal constellations that have constellation points near or at the origin, such as the signal constellation of the exemplary HSDPA signal shown in FIG. 5. With a constellation point at the origin, naturally the HSDPA has a high probability of low-magnitude events. However, if conventional hole blowing were to be applied to the signal, modulation accuracy would be substantially compromised.
Another problem with conventional hole blowing is that it ignores high-frequency events that do not have any correspondence with low-magnitude events. As shown in FIG. 6, high-frequency events 1 and 2 in the phase-difference component signal Δθ do not correlate with any low-magnitude events in the amplitude component signal ρ(t). Such a condition is seen to occur in signal constellation similar to that shown in FIG. 7, which is signal trajectory diagram of an exemplary HSUPA signal. Note that in addition to the abrupt changes in phase that occur at the origin, other high-frequency events occur between inner constellation points and peripheral constellation points. Conventional hole blowing techniques are unable to detect and correct for those high-frequency events.
Considering the foregoing limitations and deficiencies of conventional hole blowing approaches, it would be desirable to have methods and apparatus that are more effective at reducing high-frequency events in non-constant envelope polar domain signals.